Non-linear distortion generator for both second and third order distortion

ABSTRACT

An in-line distortion generator for coupling in-line with a non-linear device (NLD) produces an output signal of useful amplitude, but with low composite second order, composite triple beat and cross modulation distortions. The distortion generator comprises an instant controlled non-linear attenuator which utilizes the non-linear current flowing through a pair of diodes, in parallel with a resistor and an inductor, to provide the proper amount of signal attenuation over the entire frequency bandwidth. The distortion generator circuitry is always matched to the NLD, thereby ensuring a frequency response that is predictable and predefined. The distortion generator may also include a temperature compensation circuit to ensure consistent operation throughout a wide temperature range.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to communication systemsemploying amplification devices. More particularly, the inventionpertains to a non-linear predistortion or postdistortion generator forcoupling in-line with an optical receiver, optical laser transmitter oran amplifier to minimize second and third order distortion caused by thesignal processing.

2. Description of the Related Art

Analog intensity modulation of a distribution feedback (DFB) laser is awidely used technique to transmit analog signals, such as sound or videosignals and data, on optical fibers over a long distance. Opticaldetector also is widely used in fiber optic link. The performance of DFBlasers and optical detectors are limited by their distortionperformance. Improving second order and third order distortionperformance can greatly improve the entire system performance andincrease the entire system dynamic range.

Amplifiers are also widely used in many types of communicationapplications. Although it is preferable to keep amplifiers within theirlinear range of operation, it has been increasingly necessary to extendthe operation of amplifiers into high power and high frequency regionsof operation. Typically, the output power of an amplifier is limited bythe non-linearity of the active devices, including bipolar transistorsand FETs. These non-linearities result in distortions which areimpressed upon the signal being amplified. Reducing the non-lineardistortions of an amplifier results in increases of the output power,the system dynamic range and the carrier-to-noise ratio. Accordingly,minimizing distortions and achieving linear frequency response isparamount to efficient amplifier operation.

Minimizing distortion is particularly important when a series ofamplifiers is cascaded over a signal transmission path, such as a seriesof RF amplifiers in a CATV transmission system. Disposed throughout aCATV transmission system are RF amplifiers that periodically amplify thetransmitted signals to counteract cable attenuation and attenuationcaused by passive CATV components, such as signal splitters andequalizers. The RF amplifiers are also employed to maintain the desiredcarrier-to-noise ratio. Due to the number of RF amplifiers employed in agiven CATV transmission system, each RF amplifier must provide minimumdegradation to the transmitted signal.

Many amplifiers are subject to a wide range of ambient operatingtemperatures. These temperature changes may affect the operatingcharacteristics of certain electronic components within the amplifier,thereby inducing additional distortions. A temperature range of −40° C.to +85° C. is not uncommon for many amplifier applications in acommunication environment. To ensure consistent performance over theoperating bandwidth, and to minimize resulting distortions, an amplifiermust be designed for a broad range of ambient operating temperatures.

The distortions created by an amplifier which are of primary concern aresecond (even) and third (odd) order harmonic intermodulation anddistortions. Prior art amplifier designs have attempted to amelioratethe effects of even order distortions, such as composite second order(CSO) distortion, by employing push-pull amplifier topologies, since themaximum second order cancellation occurs when equal amplitude and 180°phase relationship is maintained over the entire bandwidth. This isachieved through equal gain in both push-pull halves by matching theoperating characteristics of the active devices. In some cases, secondorder correction is still needed in order to get good CSO performance.Many prior art designs include the use of a separate second orderdistortion circuit to provide such the correction for CSO.

However, odd-order distortion is difficult to remedy. Odd-orderdistortion characteristics of an amplifier are manifest as crossmodulation (X-mod) and composite triple beat (CTB) distortions on thesignal being amplified. X-mod occurs when the modulated contents of onechannel being transmitted interferes with and becomes part of anadjacent or non-adjacent channel. CTB results from the combination ofthree frequencies of carriers occurring in the proximity of each carriersince the carriers are typically equally spaced across the frequencybandwidth. Of the two noted distortions, CTB becomes more problematicwhen increasing the number of channels on a given CATV system. WhileX-mod distortion also increases in proportion to the number of channels,the possibility of CTB is more dramatic due to the increased number ofavailable combinations from among the total number of transmittedchannels. As the number of channels transmitted by a communicationsystem increases, or the channels reside close together, the odd-orderdistortion becomes a limiting factor of amplifier performance.

There are three basic ways of correcting distortion created by anon-linear device (NLD): 1) reduce the signal power level; 2) use a feedforward technique; and 3) use a predistortion or postdistortiontechnique. The first method reduces the signal power level such that theNLD is operating in its linear region. However, in the case of an RFamplifier this results in very high power consumption for low RFoutputpower.

The second method is the feed forward technique. Using this technique,the input signal of the main amplification circuit is sampled andcompared to the output signal to determine the difference between thesignals. From this difference, the distortion component is extracted.This distortion component is then amplified by an auxiliaryamplification circuit and combined with the output of the mainamplification circuit such that the two distortion components canceleach other. Although this improves the distortion characteristics of theamplifier, the power consumed by the auxiliary amplification circuit iscomparable to that consumed by the main amplification circuit. Thiscircuitry is also complex and very temperature sensitive.

The third method is the predistortion or postdistortion technique.Depending upon whether the compensating distortion signal is generatedbefore the non-linear device or after, the respective term predistortionor postdistortion is used. In this technique, a distortion signal equalin amplitude but opposite in phase to the distortion component generatedby the amplifier circuit is estimated and generated. This is used tocancel the distortion at the input (for predistortion) or output (forpostdistortion) of the amplifier, thereby improving the operatingcharacteristics of the amplifier.

One such distortion design, as disclosed in U.S. Pat. No. 5,703,530 andshown in FIG. 1, relies upon a traditional π-attenuation network and adelay line for gain compensation; and a diode pair coupled with a delayline for distortion and phase compensation. This circuit generates adistortion that is equal in amplitude but opposite in phase to thedistortion introduced by the amplifier. Plots of the distortionscontributed by the distortion generator and the distortions manifest bythe amplifier are shown in FIGS. 2 and 3. As shown, the distortionsignal compensates for the distortions generated by the amplifier.However, the use of delay lines in such a manner is impractical sincedelay lines are physically large, are difficult to adjust and theresults are inconsistent across a wide frequency range. Additionally,both amplitude and phase information are required for correctcompensation. The '530 patent also states that the system disclosedtherein is not ideal for certain application, such as predistortion forCATV RF amplifiers, due to the excessive losses introduced by thedistortion circuit.

An inline predistortion design, as disclosed in U.S. Pat. No. 5,798,854,provides compensation for NLDs by applying a predistorted signal equalin magnitude but opposite in phase to the distortion produced by theNLD. However, the circuitry disclosed therein is not matched to the NLD.Additionally, the '854 patent presents a design that is typical of theprior art in the use of a high resistance bias for the diodes. This willreduce the correction efficiency and increase the effects of temperatureupon the circuit.

Prior art designs also use separate correction circuits to correct forsecond and third order distortions if both types of corrections arerequired. This increases the cost of the overall circuit design and alsogenerates more circuit losses.

Accordingly, there exists a need for a simple distortion generator whichcounteracts the distortion created by an NLD. The circuit should notintroduce additional signal delay and should operate over a widefrequency bandwidth and wide ambient temperature range.

SUMMARY OF THE INVENTION

The present invention is an in-line predistortion or postdistortiongenerator for coupling in-line with an NLD to produce an output signalof useful amplitude, but with low composite second order, compositetriple beat and cross modulation distortions. The distortion generatorcomprises an instant controlled non-linear attenuator which utilizes thenon-linear current flowing through a pair of diodes to provide theproper amount of signal attenuation over the entire frequency bandwidth.The distortion generator circuitry is always matched to the NLD, therebyensuring a frequency response that is predictable and predefined. Thedistortion generator permits selective adjustment of the non-linearcurrent flowing through the diodes to create a second order distortion.The distortion generator also includes a temperature compensationcircuit to ensure consistent operation throughout a wide temperaturerange.

Accordingly, it is an object of the present invention to provide atemperature compensated distortion generator which minimizes compositesecond order, cross modulation and composite triple beat distortionsmanifested by an NLD such as an RF amplifier, a laser diode or aphotodetector.

Other objects and advantages of the of the present invention will becomeapparent to those skilled in the art after reading a detaileddescription of the preferred embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a prior art distortion generator.

FIG. 2 is a combination plot of the effect of using the outputs from theprior art distortion generator shown in FIG. 1 with an RF amplifier.

FIG. 3 is a combination plot of the effect of using the outputs from theprior art distortion generator shown in FIG. 1 with an RF amplifier.

FIG. 4 is schematic diagram of a π attenuator.

FIG. 5 is a signal diagram of the diode non-linear current caused by theinput voltage.

FIG. 6 is a schematic diagram of the preferred embodiment of the secondand third order distortion generator of the present invention.

FIG. 7 is a schematic diagram of the temperature compensation circuit.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The preferred embodiment of the present invention will be described withreference to the drawing figures where like numerals represent likeelements throughout. Although the preferred embodiment of the presentinvention will be described, for simplicity of explanation, as beingcoupled with an RF amplifier, those skilled in the art would clearlyrecognize that such a distortion generator could also be utilized tocompensate for distortion in laser transmitters, optical detectors, andother electronic components which operate over a wide range offrequencies. The description herein is not intended to be limiting,rather it is intended to be illustrative.

The present invention will be described with reference to FIG. 4,whereby a π attenuator network 20 is shown. The network 20 comprises aselected configuration of resistors Z₁, R₁, R₂, R₃, Z₀, R_(p). Thesignal source is input at signal input 30 and the output of theattenuator network 20 is seen across the output 95. Z₁ is the source ofinternal impedance which should be equal to the system impedance Z₀,which is seen across the output 95. In an embodiment of the inventionfor use with a CATV system, the impedance values Z₁ and Z₀ are equal to75 Ohms. Three of the resistors R₁, R₂, R₃ form a π attenuatorconfiguration. Preferably, the values (Y) of resistors R₂ and R₃ areequal, and substantially larger than the value (X) of resistor R₁.Resistor R_(p) is connected in parallel with resistor R₁.

As one skilled in the art would clearly recognize, when the followingcondition is satisfied:

X=2Z ₀ ² Y/(Y ² −Z ₀ ²)  Equation (1)

the attenuator network 20 is matched at input and output, from DC tovery high frequencies. For one example of the attenuator when X=7.5 andY=1.5K, the power attenuation A for this attenuator network 20 is:$\begin{matrix}{A = ( \frac{\frac{2( {{Y\quad {Z_{0}/( {Y + Z_{0}} )}} + X} )Y\quad ( {Y\quad {Z_{0}/( {Y + Z_{0}} )}} )}{( {Y + X + {Y\quad {Z_{0}/( {Y + Z_{0}} )}}} )\quad ( {X + ( {Y\quad {Z_{0}/( {Y + Z_{0}} )}} )} )}}{Z_{0} + \frac{( {{Y\quad {Z_{0}/( {Y + Z_{0}} )}} + X} )Y}{Y + X + {Y\quad {Z_{0}/( {Y + Z_{0}} )}}}} )^{2}} & {{Equation}\quad (2)}\end{matrix}$

Under the condition when Z₀<<Y, (as is the case when X=7.5 and Y=1.5K):

A≅(2Z ₀/(2Z ₀ +X))²  Equation (3)

A(dB)=10 lg A  Equation (4)

When X=7.5 and Y=1.5k, A (dB)≅0.42 dB. This means the attenuator network20 has very low insertion losses and a good frequency response. When Xhas a small variation due to the parallel of R_(p), shown in FIG. 4,from Equation (3) $\begin{matrix}{{{Delta}\quad A\quad ({dB})} \cong {{- 8.68}\quad \frac{{Delta}\quad X}{{2Z_{0}} + X}}} & {{Equation}\quad (5)} \\{{{Delta}\quad X} = {{\frac{X\quad R_{p}}{X + R_{p}} - X} = {- \frac{X^{2}}{R_{p}}}}} & {{Equation}\quad (6)}\end{matrix}$

From Equation (6): $\begin{matrix}{{{Delta}\quad A\quad ({dB})} \cong {8.68\quad \frac{X^{2}}{2Z_{0}R_{p}}}} & {{Equation}\quad (7)}\end{matrix}$

For example, If R_(p)=375 ohms then: $\begin{matrix}{{{{Delta}\quad A\quad ({dB})} \cong {8.68\quad \frac{7.5}{150}\quad \frac{7.5}{375}}} = {0.00868\quad {dB}}} & {{Equation}\quad (8)}\end{matrix}$

Equation (8) shows that when R_(p) (375 ohms) is in parallel with R₁(7.5 ohms), the attenuation will be reduced by 0.00868 dB. This amountof attenuation change is needed for non-linear compensation for anamplifier. This example also shows that when the value of R_(p)>>R₁,(i.e., when R_(p) is 50 times larger than R₁), adding R_(p) parallelwith R₁ has almost no effect on the impedance match, and the voltagedrop over the R_(p) is mainly determined by the value of R₁.

However, if a linear resistor R_(p) is used in the attenuator network20, there will be no distortion signal produced. The attenuator network20 as shown is a linear device. In order for a distortion circuit tooperate effectively, diodes are used to create a non-linear resistance.Preferably, Schottky diodes are utilized. At small current, diodecurrent is exponentially proportional to the voltage across over thediode. Thus diodes can be used as a non-linear resistance. Fornon-linear applications, the amount of attenuation can be calculated as:$\begin{matrix}{{{Delta}\quad A\quad ({dB})} = {{8.68\quad \frac{X\quad X}{2Z_{0}R_{p}}} \cong {8.68\quad \frac{X - I_{p}}{2Z_{0}I_{1}}}}} & {{Equation}\quad (9)}\end{matrix}$

Where I_(p) is the current flow through R_(p), (the non-linearresistance). I₁ is the current flow through R₁. Equation 9 provides therelationship of the attenuation change due to the current change inI_(p). This equation is accurate over a broad frequency range. Therelationship between the delta attenuation and a change in current isstill valid when the resistance is a non-linear resistor. Accordingly,Equation 9 provides a good estimation of how much non-linear current isrequired for predistortion or postdistortion purposes.

Referring to FIG. 5, when the input sinusoidal voltage wave changes fromV₁ to V₂ to V₃, the output current changes from I₁ to I₂ to I₃respectively. The non-linear current used for third order correction is:

I _(non-linear) ≅I ₁−2I ₂ +I ₃  Equation (10)

From Equation 9, the non-linear current needed is: $\begin{matrix}{{{Delta}\quad A_{{non}\text{-}{linear}\quad {correction}}\quad ({dB})} \cong {8.68\quad \frac{X\quad I_{{non}\text{-}{linear}}}{2Z_{0}I_{output}}}} & {{Equation}\quad (11)}\end{matrix}$

Only non-linear current will be useful for predistortion orpostdistortion purposes. Equation 11 can be rewritten in the form of:$\begin{matrix}{{{Delta}\quad A_{{non}\text{-}{linear}\quad {correction}}\quad ({dB})} = {8.68\quad \frac{I_{{non}\text{-}{linear}\quad {eff}}}{I_{output}}}} & {{Equation}\quad (12)} \\{I_{{non}\text{-}{linear}\quad {eff}} \cong \frac{\quad I_{{non}\text{-}{linear}}}{R_{1}/( {2Z_{0}} )}} & {{Equation}\quad (13)}\end{matrix}$

Accordingly, I_(non-linear eff) in Equation 12 is the effectivenon-linear current going to the output port 114 which is shown in FIG.6. I_(output) in Equation 12 is the total current that goes to theoutput port 114. Equation 12 also shows that it is the non-linearcurrent flowing through the diodes which causes the distortioncorrection. Any method which increases the non-linear current mayincrease the correction efficiency. Equation 13 shows that only a smallpart of the non-linear diode current is effectively being used forcorrection.

The π attenuator network 20 has low insertion loss and the voltage dropof the input voltage on R₁ (shown in FIG. 4) is proportional to theinput voltage. This voltage may be used to drive a pair of diodes toproduce non-linear current and provide third order correction. Thenon-linear current flowing in the diodes will cause an attenuator toprovide less attenuation at larger RF amplitudes, (i.e. when the inputsignal has a higher power). This may be used to compensate for thesignal compression caused by amplification. Because of the relativelyhigh value of the diode's non-linear resistance, the match of theattenuator network is almost unchanged. This match will not be changedeven over temperature. Additionally, frequency response overmulti-octave frequency bands is favorable.

The mechanisms of the second order correction circuit is also clear. Ifthe DC bias on each of the two diodes is different, for every RFpositive circle and negative circle, I_(non-linear eff) will bedifferent. Accordingly, instead of third order correction, this circuitwill also provide second order correction.

Referring to FIG. 6, the preferred embodiment of the attenuator 100 forboth second and third order predistortion and postdistortion is shown.The attenuator 100 of the present invention includes several additionalcomponents that modify a traditional π attenuator to achievesignificantly better performance over a wide frequency and temperaturerange. The attenuator 100 has an input port 101, an output port 114 andtwo bias control points 116, 123. The attenuator 100 may be used in apredistortion configuration with an amplifier or in a postdistortionconfiguration. For a predistortion configuration, the output port 114 isconnected to the input of an amplifier. For the postdistortionconfiguration as shown in FIG. 6, an output signal generated by anamplifier, is applied to the input port 101. The attenuator 100 includesresistors 105, 106, 107, 108, 112; capacitors 102, 103, 104, 111, 113,115; diodes 109, 110, and an inductor 117.

In most prior art applications, an inductor is used as a phase controlelement to change the correction signal phase. However, in the presentinvention, the inductor 117 is used in series with the resistor 108 tomake a parallel resonance circuit with the forward biased diodecapacitor. The inductive reactance cancels the specific capacitivereactance of the diodes. At the resonance frequency, the capacitance ofthe diodes 109, 110 will be compensated by the inductor 117 so that theimpedance between points 118 and 119 will be purely resistive and can becalculated as follows:

R _(impedance between 118, 119) =L/(C*R);  Equation (14)

where L is the inductance of 117 in Henrys; C is the total forwardbiased capacitor in Farads; and R is the resistance 108 in Ohms. Bycarefully controlling L and C, one may get the following:

R _(impedance between 118, 119) =R  Equation (15)

This means the capacitive effect has been totally canceled and an idealpure resistive load over a very wide frequency range has been achieved.

In prior art systems, the capacitance associated with the diodes has notbeen considered. In predistortion applications, Shottky diodes areforward biased, which results in a greater capacitance. When an RFsignal is input across the diodes, the average capacitance increases.Even at a bias of 0 volts, the impedance introduced by the diodes'capacitance may not be ignored since the capacitance in parallel withthe PN junction of the diodes will reduce the overall voltage drop onthe diodes, thus reducing the non-linear current produced by the diodesand the overall correction effect. Compensating for the capacitanceassociated with the diodes 109, 110, the inductor 117 resonates with thecapacitance of the diodes 109, 110 at higher RF frequencies, thusextending the overall frequency response of the circuit.

The function of the resistors 105, 106, 107, 108, 112 and the-capacitors102, 103, 104, 111, 113, 115 and inductance 117 is to form a modified nattenuation network in comparison to the π attenuation network 20 shownin FIG. 4. The capacitors 102, 103, 104, 111, 113, and 115 are also usedfor DC blocking and AC coupling. From an AC standpoint, the parallelcombination of resistors 105 and 106 is functionally equivalent toresistor R₂ of FIG. 4. Preferably, the values of resistors 105 and 106should be chosen such that the parallel combination is equivalent to thevalue of resistance of resistor 112, (i.e.((R₁₀₅*R₁₀₆)/(R₁₀₅+R₁₀₆))=R₁₁₂). Resistor 108 is functionally equivalentto resistor R₁ of FIG. 4; and the in-series combination of resistor 112and capacitor 111 is functionally equivalent to resistor R₃ of FIG. 4.The value of resistor 107 has no effect on RF signal attenuation.

The other function for resistors 105, 106, and 107 is to supply a majorDC bias to the diodes 109, 110. The diodes 109, 110 are first connectedin series; and the series combination is connected to resistor 107 inparallel. Because resistor 107 has a low resistance value and is inparallel with the diodes 109, 110 the voltage drop across the diodes109, 110 will be primarily determined by the resistance of resistor 107.If the DC current flow in resistor 107 is much more than the currentflow in the diodes 109, 110, the DC voltage drop across the diode 109,110, will be very stable and will be insensitive to the presence orabsence of a signal at the input port 101.

Three resistances 120, 121 and 122 act as a voltage divider to provideanother DC bias across the diodes 109, 110. As shown, resistor 121 is avariable resistor to provide a DC input bias at point 123. In thismanner, the DC bias on the two diodes 109, 110 is made unequal so thatnon-linear current produced by the positive and negative cycles of thediodes 109, 110 is different This unequal current creates second orderdistortion correction current. The correction polarity for the secondorder distortion depends upon the DC offset voltage at node 123. In thismanner, both second and third order distortion correction may beprovided.

The integrated functions of signal attenuation and diode bias supplyavoid any parasitic effects due to the introduction of additional biascircuitry. This permits a high frequency response and a favorableimpedance match.

From a DC perspective, resistor 107, in parallel with capacitors 103 and104, provides a dissipative circuit to the capacitors 103, 104.Therefore, resistor 107 will discharge the accumulated electric chargeof connected capacitors 103, 104 in every AC cycle.

Diode 109 is connected to resistor 108 through capacitor 104 while diode110 is connected to resistor 108 through capacitor 103. Diode 109 isresponsible for the RF distortion correction during the negative portionof the AC cycle, while the diode 110 has the same function during thepositive half of the AC cycle. The non-linear current of diode 109charges capacitor 104, and the non-linear current of diode 110 chargescapacitor 103. Due to the configuration of the circuit, the voltageproduced on capacitors 103 and 104 have the same value but differentsigns. The small resistance from resistor 107 connected to thecapacitors 103, 104 discharges the accumulated electric charge duringevery AC cycle. As a result, there is no additional DC voltage dropacross the capacitors 103, 104 due to the input RF signals. This permitsthe diode 109, 110 to provide the largest non-linear current for thecorrection purpose.

The present invention has several unique advantages over the prior art.This circuit provides both second and third order correction at the sametime. This makes the correction circuit very simple and effective. Theattenuator 100 uses two low series resistances 107, 108. Resistor 107significantly improves the correction efficiency and resistor 108provides for distortion correction with low insertion losses. Due to theattenuator 100 design, the voltage drop across resistor 108 fully loadsthe diodes 109, 110 even under non-linear operation of the diodes 109,110. As a result, maximum non-linear current is utilized for correctionpurposes. The present attenuator design uses low series resistance 108in series with the inductor 117 to compensate for the capacitance of thediodes 109, 110. Thus, this circuit may work over a wide frequencyrange. This correction circuit design is flexible and may be adjusted todifferent kinds of RF hybrids with different distortion characteristics.Additionally, the circuit is always matched to its input side and outputside over wide frequency range. Finally, proper phasing of thedistortion signals is inherent in the design, thereby avoidingadditional phase circuitry and delay lines. This permits a circuitdesign which is much-less complex and, therefore, is compact and robust.Table 1 provides a listing of the components shown in FIG. 6. However,one skilled in the art would clearly recognize that the values shown inTable 1 are only for explanatory purposes, and should not be consideredto be limiting to the invention. For example, the value of resistor 108may range from approximately 2Ω to 30Ω. Likewise, the value of resistor107 may range from approximately 100Ω to 3000Ω.

TABLE 1 VALUE OR COMPONENT IDENTIFICATION 102 0.1 μf 103 0.1 μf 104 0.1μf 105 6 KΩ 106 6 KΩ 107 330 Ω 108 7.5 Ω 109 HP HSMS-2822#L30 110 HPHSMS-2822#L30 111 0.1 μf 112 3 KΩ 113 0.1 μf 114 75 Ω 115 0.1 μf 117 1.5nH 120 2 KΩ 121 500 Ω 122 2 KΩ

As previously described, the attenuator 100 uses the non-linear currentproduced by the diodes 109, 110 to compensate for the second order andthird order distortion caused by an NLD. As shown, the attenuator 100comprises capacitance, resistance and two diodes. The diodes are theonly components that are sensitive to temperature change and the onlycomponents that require correction during operation over a widetemperature range. There are three factors which must be taken intoconsideration when operating the attenuator 100 over a wide temperaturerange:

1) The diode operating current will change if the bias voltage remainsconstant while the ambient temperature changes. Under the same inputvoltage swing at the input port 101 and the same bias voltage, morenon-linear diode current will be created as the ambient temperaturerises.

2) When the ambient temperature rises, the diode will produce lessnon-linear correction current for the same input signal voltage and thesame diode bias current.

3) NLDs typically exhibit more distortion as the ambient temperaturerises. Accordingly, a higher diode non-linear current is required forcorrection of the greater distortion.

All of the temperature effects experienced by the attenuator 100 arerelated to the bias voltage. Some of the effects are additive whileothers are subtractive. However, the result is that for a giventemperature, there will be an optimum bias voltage to produce the propercorrection output. Proper temperature correction will be achieved whenthere is a predefined change of bias voltage verses temperature.

Referring to FIG. 7, the preferred embodiment of the temperaturecompensation circuit 200 is shown. The temperature compensation circuit200 controls the bias of the diodes 109, 110 (shown in FIG. 6) foroptimum compensation of the distortion. As shown, the temperaturecompensation circuit 200 comprises two transistors 206, 213; a capacitor216; nine resistors 201, 202, 203, 204, 207, 209, 210, 214, 215; twodiodes 205, 208; and a negative temperature coefficient thermistor 211.

The negative temperature coefficient thermistor 211 is coupled inparallel with resistor 210 to form a temperature linearized resistance,which is correlated to a change in temperature. The PNP transistor 206provides a constant current source through its collector to thelinearized resistor combination 210,211. The constant current providedby the PNP transistor 206 induces a linearized voltage change across theresistor combination 210,211 as the temperature changes. By adjustingthe value of the variable resistor 202, the amount of constant currentthrough the PNP transistor 206 can be changed. Therefore, the voltageswing over temperature can be changed. The constant current also passesthrough the variable resistor 209, thereby creating a constant voltagedrop that is used as a starting bias point for bias voltage adjustment.By selectively adjusting the resistance of resistors 202 and 209, anycombination of voltage swing and starting bias voltage can be obtained.A NPN transistor 213, which is an emitter follower transistor, providesthe control bias voltage from line 217 through line 116 to theattenuator 100, as shown in FIG. 7. The two diodes 205 and 208 are usedto compensate for the junction voltage of the two transistors 206, 213which change over temperature.

Table 2 provides a listing of the components shown in FIG. 7. However,one skilled in the art would clearly recognize that the values shown inTable 2 are only for example, and should not be considered to belimiting to the invention.

TABLE 2 VALUE OR COMPONENT IDENTIFICATION 201 16 KΩ 202 3.3 KΩ 203 4.7KΩ 204 50 KΩ 205 1N4148 206 2N3906 207 2 KΩ 208 1N4148 209 1.5 KΩ 210 2KΩ 211 DKE 402N10 212 100 Ω 213 2N3904 214 100 Ω 215 3 KΩ 216 50 μƒ

It should be recognized that the present invention provides an instantvoltage controlled non-linear attenuator design combined with a biassupply for optimum non-linear correction efficiency and bias temperaturestability. Even if the temperature compensation circuit 200 as disclosedherein is not utilized, the preferred embodiment of the presentinvention provides adequate distortion correction over a broadtemperature range. When the temperature compensation circuit 200 isutilized, the distortion compensation results can be further improved.Accordingly, a trade off between the performance of the compensatingcircuit and the complexity of the circuit must be weighted.

What is claimed is:
 1. An external distortion control circuit forselective attenuation of a CATV signal comprising: a signal input port(101); a non-linear circuit coupled to said input port and comprising: amodified π attenuator network comprising first and second resistors(105, 106) coupled in parallel, said resistors coupled in series with athird resistor (108), an inductor (117) and a fourth resistor (112);first and second diodes (109, 110) each coupled in parallel with saidthird resistor (108) and said inductor (117); first and secondcapacitors (103, 104) coupled in parallel with said diodes (109, 110); afifth resistor (107) coupled across both first and second diodes (109,110) and coupled across both first and second capacitors (103, 104), fordissipating accumulated charge on said first and second capacitors (103,104) thereby allowing full nonlinear performance by said diodes (109,110); a voltage divide (120, 121, 122) coupled with said first resistor,for adjusting a first voltage bias separately across each of said diodes(109, 110); and an output port (114) for outputting said selectivelyattenuated signal from said non-linear circuit; whereby said first,second and fifth resistors (105, 106, 107) provide a second bias voltageacross said diodes (109, 110).
 2. The distortion control circuit ofclaim 1 further including a temperature compensation circuit coupledwith said voltage divider, for selectively adjusting said DC biasvoltage in response to a change in ambient temperature.
 3. Thedistortion control circuit of claim 1 wherein said third resistor (108)and said inductor (117) generate a voltage proportional to said inputsignal; whereby said proportional voltage creates a non-linear currentthrough at least one of said diodes in said pair (109, 110), therebycreating a non-linear resistance to selectively attenuate said signal.4. The distortion control circuit of claim 2 wherein said temperaturecompensation circuit comprises: a constant current source transistor(206); a second transistor (213), coupled to the output of said currentsource transistor (206), for outputting said DC bias voltage; alinearized resistance circuit having a thermistor (211) coupled inparallel to a second resistor (210); and a variable resistor (209) thatcouples said current source transistor (206) to said linearizedresistance circuit; whereby the linearized resistance circuit iscorrelated to a change in ambient temperature.
 5. The distortion controlcircuit of claim 1 whereby said non-linear circuit provides selectiveattenuation of the signal based upon the signal magnitude; whereby lessattenuation is provided for larger signal magnitudes and moreattenuation is provided for smaller signal magnitudes.
 6. The distortioncontrol circuit of claim 1 wherein the insertion loss at said outputport is less than 0.5 dB.